Guided wave frequency range transducer



Jan. 27, 1953 J. R.`P|ERcE 2,626,990

GUIDED WAVE FREQUENCY RANGE TRANSDUCR Filed May 4, 1948 10 Sheets-Sheet l ATTORNEY Jan. 27, 1953 J. R. PIERCE 2,626,990

GUIDED WAVE FREQUENCY RANGE TRANSDUCER Filed May 4, 1948 l0 Sheets-Sheet 2 A ORNEV Jan. 27, 1953 J. R. PIERCE 2,526,990

GUIDED WAVE FREQUENCY RANGE TRANSDUCER Filed May 4, 1948 10 Sheets-Sheet 3 E94 FIG. 8 TRANSl/CER l n x l Pur/ lH6 I//4 (//6 (//6 GUIDE A ORA/EV Jan. 27, J. R. PIERCE 2,626,990

GUIDED WAVE ,FREQUENCY RANGE 'RANSDUCER Filed May 4, 1948 l0 Sheets-Shes?, 4

ATTORNEY Jan. 27, 1953 J. R. PIERCE 2,626,990

GUIDED WAVE FREQUENCY RANGE TEANSDUCER Filed May 4, 1948 l0 Sheets-Sheet 5 F IG. 24

A T TOR/VE V J. R. PIERCE GUIDED WAVE FREQUENCY RANGE TEANSDUCEE Jan. 27, 1953 l0 Sheets-Sheet 6 Filed May 4, 1948 /NVENTOR y J R. PIERCE Jan. 27, 1953 J. R. PIERCE GUIDED WAVE FREQUENCY RANGE TRANSDUCER iled May 4. 1948 l0 Sheets-Sheet 8 meal/mcy M. c/s.

Jan. 27, 1953 J. R. PIERCE GUIDED WAVE FREQUENCY RANGE TRANSDUCER Filed May 4, 1948 l0 Sheetg-Sheer. 9

0/5 TA NC E C 8 6 D" /20 0 3 4%5 5 M m m m f aww W 5 MS B58 ,6 www w m m F 5 w m 55 lim 51|, A l M 41|r1|rwlt5 7,00 d f .ad L 11(5 A?. 5 76 M www m 4 |V.||\\|\ a. L 5 F T/W l w 5 /w 2 5 To, E

' J. R. P/fRcE Jan. 27, 1953 .1. R. PIERCE GUIDED WAVE FREQUENCY RANGE TRANSDCER 10 Sheets-Sheet 10 Filed May 4. 1948 oo L v H6143 /N VE N TOR HVJ. R. P/Epcg FREQUENCY--- F Patented Jan. 27, i953 GUIDED WAVE FREQUENCY RANGE 'ERANSDUCERk .lohn R. Pierce, Miilburn, N. J., assigner to Bell Telephone Laboratories, Incorporated, New York, N. Y., a corporation of New York Application may e, 1943, Serial No. 25,027

41 Claims. l

This invention relates to novel structures adapted primarily for use at very high, ultrahigh and super-high frequencies, i. e., at fre,- quencies within the approximate ranges of 30 to 300 megacycles; 300 to 3,000 megacycles and 3,000 -megacycles and above, respectively; as fre quency selective or phase and/or amplitude equalizing devices.

Since, with frequencies within the abovementioned frequency regions, wave guide and/or coaxial line structures are usually employed for transmission purposes, and the coaxial line is, in a sense, a special form of Wave guide, we shall, throughout this application and the appended claims, refer to the over-all frequency range of interest, i. e., frequencies of approximately 30 megacycles and above, as the guided wave frequency range.

In general, because at least one cross-sectional dimension of a wave guide of the single tubular conductor type should be approximately one half a wavelength, or more, of the lowest frequency to be transmitted through it, Wave guides of this type are, as a practical matter, usually employed only at frequencies suiiiciently high that the maximum cross-sectional wave guide dimension required is not greater than about 3 inches. This virtually means that, as a practical matter, wave guide structures Will usually be employed only when the lowest frequency to be transmitted is not less than about 2,000 megacycles, corresponding to about 15 centimeters wavelength. Much larger wave guides, and correspondingly lower frequency operation can, of course, be employed if only relatively short wave guides are required or if other considerations outweigh the cost of the larger size of wave guide. At lower frequencies than 2,000 megacycles coaxial structures will, however, ordinarily be used. Coaxial structures can, of course, be used at frequencies of several times 2,000 megacycles so that there is an appreciabley frequency range within which either coaxial or wave guide structures or combinations of the two types can be employed.

For a discussion of wave guides of the single conductor type, reference may be had to the Radio Engineers Handbook by F. E. Terman, pages 251 to 264, published in 1943 by McGraw- Hill Book Co., Inc., New York city. Unless otherwise specified the term wave guide will be used throughout this specication and the appended claims to denote structures of the type described by Termen. .In general, the resonators ,employed in the numerous structures illustrative of resonators.

the` present invention, as shown in the accompanying drawings and described in detail below, are known to those skilled in the art as cavity For example, for a discussion of cavity resonators, see pages 143 to 148, inclusive, of F. E. Termans book entitled Radio En gineering, published in 1947 by McGraw-.Hill Book Co. Inc., New York city.

More particularly this invention relates to novel structures which can be employed in the guided wave frequency range to provide characteristics closely approximating those of the so-,called lumped-element lower frequency devices, commonly designated as wave filters or phase, and/or amplitude, equalizing networks, at voice and the lower carrier telephone frequency ranges.

For the lumpedvelement low frequency devices and for guided wave frequency range Wave filter and equalizing devices, it has heretofore been the almost universal practice to interconnect the input and output circuits from which and to which power is to be transferred, respectively, by a plurality of resonant circuit structures coupled in tandem, so that all portions of a signal, in passing from the input circuit to the output circuit, pass through all of the structures.

In the case of the structures of the present invention, the input and output circuits (usually Wave guides or coaxial lines) are connected by each of a plurality of resonating structures or resonators so that each resonator acts, to some degree at least, as an independent coupling and the individual structures can thus be regarded as being in parallel relation with respect to each other in the coupling afforded by their combined action between the input and output circuits. For this reason the structures of the invention can be conveniently designated generally as paralleled resonator transducers.

The term transducer for the purposes of this specication and the appended claims, .sha-l1 be understood to specically include very high, ultra-high and super-high frequency structures having characteristics similar to low frequency tuned circuits, wave filters, amplitude equalizers, phase equalizers and delay networks or delay equalizers and to mean generally any combination of structure a function of which is to transfer some portion, or all, of the energy from one circuit to another, i. e., any device of the specific or general classes named is, for the purposes of this application, a transducen A rst subclass cf structures of the invention comprises those in which a desired transmission characteristic is achieved simply by totalizing the virtually independent contributions of a plurality of resonators. In this subclass of structure the resonant frequencies of the component resonators will lie within the frequency band transmitted by the over-all structure. This subclass of structures is the guided wave frequency range counterpart of the low frequency tuned circuit type of frequency selective network in which two or more resonant combinations of a coil and a condenser each, are combined to produce relatively simple frequency selective characteristics. Such elementary circuits are used in the low frequency art, for example, in coupling the successive stages of a vacuum tube amplifier and for similar purposes where the phase and attenuation characteristics need not rigorously meet preoise and severe requirements.

Since some of the individual resonant structures can, as will become apparent hereinunder, be arranged, in any one of a number of ways, to introduce a change of phase, in transmitting energy from the input circuit to the output circuit, which differs by 180 degrees from the change of phase introduced by others of the resonant structures, a second subclass of structures of the invention can be distinguished, which second subclass includes structures having two or more resonators one portion of which resonators provides what will be designated as ldirect coupling, i. e., a coupling providing no significant phase change, and another portion of which resonators provides what will be designated as contrary coupling, i. e., a coupling which introduces a phase change of 180 degrees with respect to that of the first-mentioned portion, in transmitting electrical energy from the input to the output circuit. As will become apparent hereinunder, this second subclass of structures can readily be proportioned so as to closely simulate by the combined action of the direct and contrary coupled resonators at very high, ultra-high and super-high frequencies, the transmission characteristics of the well-known "lumped-elemen low frequency lattice type of structure. In this second subclass of structure, a number of the resonators can, as in the low frequency lattice type of structure, have resonant frequencies lying outside the frequency band transmitted by the over-al1 structure. be explained in detail hereinafter.

The lattice type structure is, of course, recognized as the most general type of lumped-element electrical network. Highly developed and refined theories and design methods for appropriately proportioning the component lumped elements and arranging such structures to provide almost any desired transmission characteristics at lower frequencies have been available to those skilled in the art for many years. For example, see the papers entitled A General Theory of Electric Wave Filters by H. W. Bode, published in the Massachusetts Institute of Technology Journal of Mathematics and Physics, No-

The significance of this will 'f vember 1934 and Ideal Wave Filters by H. W.

Bode and R.. L. Dietzold published in. the Bell System Technical Journal, volume XIV, No. 2, April 1935 at page 215.

Broadly stated, the object of the invention is to provide novel types of very high frequency, ultra-high frequency and super-high frequency (i. e. guided wave frequency range) transducers.

A principal object of the invention is to provide novel structures for use in guided wave frequency range transmission systems, which will have, in said frequency range, substantially the characteristics possessed at low frequencies by lumped-element electrical wave filters.

Another object is to provide novel structures for use in guided wave frequency range transmission systems, which will have, in said frequency range, substantially the characteristics, possessed at low frequencies, by lumped-elemen electrical equalizing networks.

A further object is to provide novel structures, for use at very high, ultra-high and super-high frequencies, which will closely simulate the characteristics possessed at low frequencies by lumped-element electrical lattice networks.

Other and further objects will become apparent during the course of the following description and from the appended claims.

The principles of the invention will be more readily understood from the following detailed description of specific illustrative embodiments of the invention and from the accompanying drawings in which:

Fig. 1 shows one form of structure of the invention which can be proportioned to be an ultra-high frequency or super-high frequency, selective wave filter, or an equalizing network, of the shunt coupling type;

Fig. 2 shows a second form of structure of the general type shown in Fig. 1 but rearranged slightly to permit somewhat improved performance;

Fig. 3 shows a structure of the invention similar to that of Fig, 2 but of the series coupling type;

Fig. 4 shows a branching lter which includes four wave filter structures of the general type illustrated in Figs. 1 and 2, assembled to connect one wave guide to four other wave guides so that four frequency bands, B1 to B4, inclusive, present in the first-stated wave guide, can be separated and directed, one each, into the four other wave guides, respectively;

Figs. 5, 6, 7, 8 and 9, inclusive, are diagrammatic showings employed in explaining the nature and functioning of devices of the invention of the general types illustrated in Figs. l to 4, inclusive;

Fig. 10 shows a two-channel branching wave filter combination of the invention which includes an impedance corrective resonator arranged to improve the impedance of the structure between the frequency bands segregated by the wave filters;

Figs. 11A, 11B and l2 show top, side and end views respectively of another structure of the invention;

Fig. 13 shows a cross-sectional view through two resonators and the wave guides of the structure of Figs. 11A, 11B and 12;

Figs. 14 to 19, inclusive, show various forms of resonator and coupling arrangements which can be employed in the structures of the invention;

Figs. 20 to 23, inclusive, illustrate a further variation in structures of the invention whereby direct and contrary couplings are effected by the positioning of the resonators;

Fig. 24 shows an arrangement of the invention in which the coupling of the resonator to the wave guides is effected through irises;

Figs. 25 to 28, inclusive, illustrate structures of the invention in which the closed ends of the two waveguides coupled thereby are Yoffset by a quarter wavelength;

Figs. 29 to 33, inclusive, illustrate alternative forms .which arrangements ofthe invention may take;

Figs. 34 to 36 are schematic electrical circuit diagrams employed in explaining the nature and functioning of the arrangement of Fig. 33;

Fig. 37 shows an illustrative model device constructed in accordance with the principles ofthe invention;

Fig. 38 shows the measured frequency-versusloss characteristic of the device of Fig. 37;

Fig. 39A shows, in cross-section, the detailed arangement of one of the resonators employed in the device of Fig. 37;

Fig. 39B is a curve of the voltage distribution along the resonator of Fig. 39A;

Fig. 40 shows a branching wave filter arrangement of the invention, employing, essentially, two devices of the type illustrated in Fig. 37, connected to a single input wave guide to separate two frequency channels or bands, and direct each band into a different one of the two output wave guides;

Figs. 41A. to 41C, inclusive, show the structure of a transducer of the invention suitable for coupling two coaxial transmission lines, the transducer being useful at very high and ultrahigh frequencies; and

Figs. 42 and 43 are admittance diagrams ernployed in, explaining further capabilities of transducer structures of the invention.

In view of the current interest in the so-called microwave radio transmission systems, in which frequencies of several thousand megacycles are commonly employed, the majority of the structures employed for illustrative purposes in this application are of types more directly suitable for use with microwave transmission systems. Similar or analogous structures, suitable for use at somewhat lower frequencies with coaxial lines can, however, be readily devised, by those skilled in the art, by the obvious application and adaptation of the principles cf the invention to corresponding coaxial line structures. Figs. 41A to 41C, inclusive, represent in detail one such adaptation. Throughout this application and the appended claims microwave frequencies shall be understood to mean those frequencies which are conveniently transmitted through the hollow tube type of wave guide of practicable physical dimensions, i. e., frequencies of approximately 2,000 megacycles and higher.

In more detail, in Fig. 1, are shown two wave guides I and 2, wave guide I being designated as the input and wave guide 2 as the output, for convenience in describing the arrangement only. The right end of wave guide i and the left end of wave guide 2v are closed. rlhe functions of these guides can usually be interchanged, if desired', without appreciably changing the ultimate result, for the accomplishment of which the arrangement has been devised. The suitable proportions for wave guides to be used in transmission systems, as is well known to those skilled in the art. depend upon the frequencies to be ernployed. The matter is discussed above and also, in some detail, in the copending application of W. D. Lewis Serial No. 789,985, filed December 5, 1947 which matured into United States Patent 2,531,447, granted November 28, 1950.

Connecting wave guide l with wave guide 2 are five resonators 3 to l,k inclusive, each being provided with a tuning stub 3. The number of resonators to. be used in any instance is determined by the Width of the. band of frequencies to be. transmitted from one wave guide to the other and by the degree ofA uniformity of transmission desired throughout that band. The wider the frequency band and the greater the uniformity desired the larger, in general, will be .the number ofresonators which should beemployed. In some instances the number of resonators required canbe decreased by using more complex types of resonators. such, for example, as those disclosed inY UnitedV States Patent 2,432,093 granted December 9. 194.7, to A. G. Fox.

Five resonators 3 to 7, inclusive, are shown in Figs. 1 and 2 for illustrative purposes, buty a greater or lesser number can equally well be employed depending upon factors, some of which havev been mentioned above 4and others of which will be discussed hereinunder.

The tuning stubs 8 can preferably be threaded and the holesA through which they. project into their respective resonators can also preferably be threaded to t the stub threads so that the stubs .can readily be adjusted with respect to the distances to which they extend into the resonators.

Each resonator is tuned to a different resonant frequency. For structures falling within the first subclass mentioned above all the resonant frequencies will lie within the band of frequencies to be transmitted by their combined` action. For structures falling within the second subclass mentioned above, some of the resonant frequencies can be located outside the band of frequencies to 'oe transmitted. In general, the resonant frequencies of a group of resonators such as 3 to l, inclusive, will usually be arranged so that, for example, the resonator 3 is tuned to the lowest resonant frequency, resonator 4 to the next. to lowest resonant frequency, and so on, resonator 'l being tuned to the highest resonant frequency. Such an arrangement of the frequencies facilitates the straightforward tuning and adjusting of the over-all transducer structure. However, it is not essential to follow such an arrangement and in cases where objectionable interference or cross-talk between adjacent resonators might be encountered if their respective resonant frequencies are too close together in frequency, or are harmonically related, the difculty can frequently be substantially reduced or eliminated by an arrangement in such order that resonators which tend to cause mutual interference are widely separated from each other.

The resonators 3 to 1, inclusive, can be substantially empty cavities enclosed by a conductive sheet 0f material. Alternatively, they caninclude, within their respective cavities, probes, or rods, or the like, arranged in conjunction with the cavity to effect resonance in man-y ways, well known to those skilled in the art, some illustrative formsM of which will be described in detail below. The resonators can be of rectangular, circular. ovoid or of substantially any convenient crosssectional and longitudinal oonformations which will provide a cavity of suitable size to afford adjustment of its resonances over the desired range by a convenient tuning arrangement. A number of illustrative forms of resonators, suitable for use in arrangements of the invention, will be de scribed below, in detail, in connection with other figures of the drawings. In general, the resonators should, for convenient use, in arrange,- ments of the general type shown in. Fig.. 1, have at least one dimension in common so that they can conveniently be assembled adjacent to the two wave guides to be coupled. Very often .the resonators can be, simply, sections of transmission line of like kind and cross-sectional dimensions with the transmission lines being coupled together by the resonators, the length of resonators then being approximately one-half wavelength of the median frequency of the band of frequencies to be transmitted.

Each resonator is coupled between the two Wave guides, the point of coupling of a particular resonator to one guide being separated appreciably (usually by approximately one-half wavelength of the median frequency of the band to be transmitted) from the point of coupling the same resonator to the other guide. Numerous and varied types of suitable couplings between the individual resonators and the wave guides are also described below, for purposes of illustration, in connection with other figures of the drawings. In some instances the coupling arrangements, particularly, for example, where iris coupling arrangements are employed, can themselves introduce reactive effects which in cooperative relation with the resonance of the main cavity determine jointly the transmission characteristics of the over-all device. For example, see the abovementioned patent of A. G. Fox in which resonant irises, many forms of which are well known to those skilled in the art, used as coupling means, cooperate with resonant cavities in determining the transmitting characteristics of his over-all arrangements.

The resonators are spaced 'along the wave guides in Fig. 1, from center to center of successive resonator to wave guide coupling points, at approximately one-half wavelength intervals of the median frequency of the channel or band of frequencies being transmitted. The resonators nearest a closed end of a wave guide, in Figs. 1 and 2, are spaced so that the coupling point centers are substantially one-quarter wavelength of the median frequency of the frequency range of particular interest, from the end of the wave guide, in each instance. The other resonators are preferably spacedso that the coupling point center of each resonator is one-quarter wavelength plus an integral number of half wavelengths, of the median frequency of the band of frequencies transmitted through it, from the closed end of the wave guide to which it is coupled.

It is assumed, in Figs. l to 4, inclusive, as will usually be the case in practice, that the coupling points between resonators and wave guides lie on the longitudinal center lines of the resonators.

Throughout this specification, its accompanying drawings, and in the appended claims, when wavelength is referred to in connection with wave guide structures, it is to be understood,`un less otherwise specified, that the wavelength referred to is the wavelength within the structure, rather than the wavelength of the same energy in free space. As is well known to those skilled in the art, the wavelength of a particular frequency of microwave energy in free space is different from its wavelength when being transmitted through a wave guide structure. In general, the wavelength of energy of a particular frequency in a wave guide is usually greater than for the same frequency of energy being transmitted through free space.

The arrangement of Fig. 2 is substantially identical with that of Fig. 1 except that the two wave guides I and 2 are both closed at their right ends and open at their left ends as shown. Their right ends are in vertical alignment. This arrangement permits each of the resonators 3 to 1, inclusive, to be more readily spaced from the closed ends of both the Wave guides so that its longitudinal center line (which should preferably also be an axis of symmetry and should include the center line of the resonator to guide coupling arrangements) is more precisely an odd number of quarter wavelengths of the median frequency of the specific band, channel, or group of frequencies, to be transmitted through the particular resonator in passing from one wave guide to the other. As contrasted with this, in Fig. 1, some compromises in spacing will be required since each resonator should be positioned, as precisely as practicable, an odd number of quarter wavelengths of the median frequency of its particular pass-band from the closed ends of both the wave guides. The preferred spacing is expressed by the relation where n is any whole number and A is the wavelength (in the guide) of the median frequency of the band passed by a particular resonator the position of which is to be determined.

In both Figs. 1 and 2, the resonators are coupled to the wave guides in what is designated in the art as a shunt relation. The meaning of the terms shunt and series in connection with wave guide couplings, or junctions, is discussed in detail in the above-mentioned copending application of W. D. Lewis. The newly published text Microwave Mixers by R. V. Pound, McGraw-Hill Book Co., Inc., New York city, 1948, at pages 259 to 262, also explains these terms as applied to wave guide junctions. The discussion below, in connection with Figs. 6 and 7, will tend to further clarify the meanings of these terms, from an electrical circuit standpoint, as employed in this specification.

In Fig. 3 the two wave guides II and I2 are coupled to opposite ends of the resonators I3 to I6, respectively, in what is designated in the art as a series relation.

A tuning stub II is provided in each of the resonators I3 to I6, inclusive, to permit suitable adjustment of their respective resonances. The center line of each of these resonators, that is, their respective points of coupling to the wave guides, should be substantially a whole number of half wavelengths from the ends of the wave guides as expressed by the relation where n is `any whole number.

In many cases it will be possible to select any one of the three forms, illustrated in Figs. l to 3, inclusive, to provide the desired transducer action. In general, the choice between the shunt type of Figs. 1 and 2 or of the series type of Fig. 3 will be determined by the ease of coupling the particular form or shape of resonator used, to the particular form or shape of Wave guide used (rectangular, square, or round, etc).

In Fig. 4 an adaptation and extension of the arrangement shown in Fig. 2 is illustrated which will provide for the separation of four frequency bands or channels, designated, for example, Bi, B2, B3 and B4. These channels could be, for eX- ample, four of the ve channels of the illustrative microwave system described in detail in the above-mentioned application of W. D. Lewis.

The resonators 26 to 31, inclusive, are apportioned, three to a filtering structure, as shown in Fig. 4., each group of three being adjusted so as to pass or transmit its particular band or channel of frequencies only, to its associated channel or branching Wave guide of vthe wave guides 22 to 25, inclusive, respectively, from the main wave guide 2|, in which all of the channels B1 to B4, inclusive, are present.

The center line of each resonator is an odd number of quarter wavelengths from the closed (right) end of main wave guide 2 as Well as from the closed end of its associated 4branching wave guide. In each instance the wavelength is preferably taken as that of the median frequency of the group o f frequencies passed, or transmitted, by the particular resonator being considered.

In Fig. 5 a block diagram representation of a conventional lumped-element lattice structure is shown in which transformer 4i] represents an ideal transformer providing whatever impedance transformation (if any) is effected between the input terminals 43 andthe output terminals 44 of the structure. In many cases no impedance transformation is desired, in which cases the ideal transformer should have a 1:1 ratio. The lattice structure proper is represented by the series arm admittances 4I, each being designated and the shunt, or crossed, arm admittances 42, each being designated and the other of which provides a contrary coupling having a total admittance of In Fig. 6 an electrical schematic diagram of a circuit is shown, which `is substantially a low frequency or lumped-element equivalent of the general type of the devices of Figs. ll and 2 and which can be made equivalent to-the lattice structure of Fig. 5, provided that admittances is Yand Y are suitably proportioned. In this figure, which is a reasonably accurate representation of the structures of Figs. 1 and 2, if the resonators are substantially loss-free,

can be represented by a resonant circuit or a plurality of resonant circuits such as those comprising inductanoe 5S and capacity 57 connected in series and inductance 5S and capacity 59 connected in series, the transformers 51%, lill, 55 and 6I serving to furnish any impedance transformations that may be required' to provide matched impedances throughout the circuits and to couple their respective input and output ends to the input terminals 5i), 5l vthrough the associated conductors 59, 5l and to the output terminals 52, 53 through the associated conductors a2', 53', respectively, as shown in Fig. 6.

In like manner can be represented by one or more resonant circuits such as those comprising inductance 64 and capacity $5 connected in series and inductance 66 and capacity 61 connected in series, the transformers G2, 58, 63 and 69 serving to furnish appropriate impedance transformations, if required, and to couple their respective input and output ends to the terminals 5,9, 5| through the conductors Et', 5| and to the terminals F22, 53 through the conductors 52V', 53', respectively, as shown in Fig. 6.

One of the four pairs of input or output transformers of Fig. 6, i. e., one of the pairs of transformers 54, 55; 62, 63; 6B, iii; or 63, 69; should be connected to provide a reversed coupling to input terminals El), 5l or to output terminals 52, 53.

In order to correspond diagrammaticaliy more closely to general concept as represented by the lattice network of Fig. 5, it might be preferable to have transformers 54, 55 or 60 and 5| provide the reversed or contrary coupling, though the end result is, obviously, the same if the reversed coupling is provided by transformers 62, 53 or G8 and 59. As shown in Fig. 6, transformers 62, 53 are reversedy with respect to transformers Ed, in coupling to the input terminals lill, 5I.

The breaks represented by dashed lines fit, in the conductors 56', Y5I 5,2', 53 are to indicate that other direct and contrary coupled resonant circuits can be inserted, so as to make a trans.1 ducer with more than four resonators.

The lower ends of conductors Ell', lil' and 52', 53 are left open-circuited to correspond to the high impedance which is present at the coupling points, by way of example, of resonator 3 of Fig. -2 and results, of course, from the quarter wavelength spacing from the short-circuited ends of Wave guides I and 2.

All of these transformers are usually simple unit ratio substantially ideal transformers, though an appropriate impedance .transformation could, obviously, readily be introduced by the transformers were the impedance of the source connected to the input terminals different from that of the load connected to the output terminals. The impedance level of the coupling circuit can obviously also be raised or lowered to afford more conveniently realizable inductanoe or capacity values or more convenient dimensions for the resonators in high frequency structures, and appropriate impedance transformations to match the yimpedance of the source and load can then be introduced by the coupling transformers, or their counterparts, i. e. `the high frequency coupling structures. The most efficient transfer of power requires, of course, the matching of the impedanceswat all junction points of the system.

1i r'Obviously also, any desired complexity of the admittances YorY can be realized by utilizing an appropriate nurnber of resonant circuits, which could appear between dashed lines 48 and 49 and below line 47 in Fig. 6, so that it is apparent that a structure of the general character represented by the schematic circuit of Fig. 6 can accurately simulate any lattice type structure, as illustrated in Fig. 5. Since all of the resonant circuits are connected in shunt across the input and output terminals 50, I and 52, 53, this type of structure is designated as being of the shunt type. The terminals are, of course, the low frequency counterparts (or substantially equivalents) of the input and output ends of the wave guides I and 2 of Figs. 1 and 2, respectively, the resonant circuits are, speaking generally, the low frequency counterparts of resonators such as the resonators 3 to 1, inclusive, of Figs. 1 and 2, and the transformers are the low frequency counterparts of the coupling arrangements connecting the wave guides and resonators including the eiects of the spacing of the resonators relative to one another and to the closed ends of the wave guides.

The alternative series type of circuit, shown in microwave" form in Fig. 3, is illustrated by the equivalent low frequency electrical schematic diagram of Fig. '7. Since the series type of circuit is more conveniently manipulated mathematically on the basis of impedance, direct and contrary coupled impedances s. m Z and Z are indicated.

The impedance is represented as comprising two parallel resonant combinations (sometimes called antiresonant combina-tions) comprising coil 80 and condenser 8| connected in parallel, and coil 82 .and condenser 83 connected in parallel, with input transformers 'I6 and 'I1 and output trans- .formers 'I8 and I9 coupling these combinations to the input terminals 10, 'Il through conductors and II and the output terminals 13, 'I4 through conductors 'I3' and 14', respectively.

Similarly, the contrary connected impedance is represented by parallel resonant circuits com- .prising inductance 88 and capacity 89, connected in parallel, and inductance S0 and capacity 19|, .connected in parallel, the circuits in turn being coupled to terminals 10, 'II and 13, 'I4 by conductors 10', II and 13', 'I4' respectively and by input transformers 84, 85 and output translformers 86, 81, respectively, the last-stated transformers providing, in this instance, the contrary coupling desired.

AS in the case of Fig. 6, the transformers of Fig. 7 are normally of unity ratio but can be .proportioned to introduce an appropriate change in impedance should the source be of different impedance from the load or should it be desirable .to raise or lower the impedance level of the resonant combinations to obtain Amore favorable or convenient impedance element values.

In the case illustrated in Fig. '7, thetrans- `formers are connected in series between the input and output terminals, and the local (lower) ends of the conductors 10', 'II' and 'I3'. 'I4' are shortfcircuited by conducting members 'I2 and 15, respectively. This corresponds, as is well known to those Skilled in the art, to the eiective short .circuit appearing at the point of coupling of resonator I3 of Fig. 3 Vby virtue of the half wavelength spacing from the closed ends of wave guides II and I2. The breaks represented by ldashed lines 48, 49 indicate where further resonant circuits and transformers may be connected.

As in the case of the circuit of Fig. 6, in Fig 7 one resonant circuit or more than two resonant circuits can be employed to provide the desired impedances i-I- Z or Z as convenience in simulating the required im- .pedances may dictate, two having been shown in both Figs. 6 and 'I solely for purposes of illustration.

In both Figs. 6 and 7 the couplings provided by all of the transformers are normally substantially identical except for the inversions required .to provide contrary couplings. Increasing the coupling of any transformer, however, is equivalent to `decreasing the inductance and increasing the capacity of the associated resonant circuit. Alternatively a single transformer can obviously be employed to couple two or more resonant combinations to the terminals, where no unique change in impedance or direction of coupling is required for any individual resonant combination.

Tuning a resonator, as for example, by adjusting the protrusion of the tuning stub in a resonator used in the devices of the invention, such as those shown in Figs. 1 Ito 3, inclusive, has the effect of changing the effective inductance or capacity or both of the resonator.

While the potential equivalence of the arrangements of the invention, whereby one wave guide is coupled to another through a plurality of .microwave resonators, some of which provide direct and others of which provide contrary coupling, to the lattice type network of the lumped-element, low frequency kind, so well known to those skilled in the art, may appear `evident from the above description of Figs. 5, 6 land 7, it can be made more clearly apparent by the following simple analysis of one phase of this relation.

The arrangement of Fig. 6, by way .of example, can be represented by the simplied block schematic of Fig. 8, where all the resonators taken collectively can be represented by block 92 as a four-terminal transducer. Admittances S3 and l94 represent the characteristic, purely conductive admittance (or conductance) M of appropriate wave guides, which are assumed to be connected to the input and output terminals of the transducer 82, as shown. These admittances are assumed to be substantially pure conductances, as will be the case for properly proportioned and terminated wave guides.

The energy of the source is represented by an impressed current of 2, in accordance with a conventional well-known method of network analysis. This signifies that I is the current Iwhich iiows in the output termination 94 of Fig. 8 when transducer 92 is removed and the load and source are directly connected.

. 13 lThe following equations will then be applicable:

we may "Seite these for the' input .admittance or the transducer andreas; Y1, ane the raue we 1f Infidel, the tierischen-'gives perfect trainsmission.

NOW,

Ll. 2. Y and Y areadmittanee functionsas illustrated by lcurves 95and 96', respectively, ffi'g., '9,11avin`g a n v.mber of poles orantire'sonances, vttc h, 'inc slve, one located at they resonance of each' resonator, and "zeros, lor resonances, i to1i,lin'clu"sive,be tween the poles Since1'the'admittancefYWis the reciprocal of theiim active structure the e's nancesfiofthe---admittance curves will correspond'infrequency'with antiresonances "of the impedance-curve and antiresonance of the "admittance curve 4will correspond in frequency' .with resonances of the impedance curve. lIn either cased-an' 'antiresonl ance must voccur between Atwo #successive resonance musi'I occur between two successive' resonances and vice l versa. See A p Reagctance Theor'um. y-b'y V`Ronald A. Foster-Bell System Technical'Jornahvolume 3; page 25,9, npriljl924. Accordingly theadmittance :curves of Fig. 9 correspond toa transducer of the invention having a total of 'eight' resonators, four of :which `are direct coupled andV four of whichare contrary coupled. The adrriittance` of`*the yfinir direct coupled resonators'i's'thenepresen'tedby curve 95and these four resonatoi'sare resonant atfr'equencies ia, b; c, and d, respectively. Similarly, the admittance `of the four contarywcpled resonators is represented by'curve .fii'al'fld 'theffor contrary 'coupled` resonators are resonant atfrequencies fe, i; gand'h, respectively. "'Severalforms of Ytransducers lof the inventionhaving-eight resonators, four of which can be direct coupled` and four of which can be contrary coupled are described in detail hereinunder.v Since in 'Fig'. 9 the resonant and antiresonant frequencies, z', y, lc, and b, c, d, respectively, of curve correspond in frequency with the antiresonant and resonant frequencies, e, f, y, and l, m', n, respectively, of curve 96 and antiresonances a and h of curves 95 and 96, respectively, have no corresponding critical frequency (pole or zero) inthe other curve, by elementary lattice type lter theory, the transducer having its resonators adjusted to produce admittances corresponding t0 those represented by curves 95 and 96 of Fig. 9, is a bandpass wave filter having itsflower cut-off at frequency a, its upper cut-off at frequency h and transmitting the band of frequencies between frequencies a and h. Atransducer of the invention having two groups or sets of resonators, one of which comprises direct coupled resonators and the other of which comprises contrary coupled resonators, the resonant frequencies of which resonators, are arranged generally as illustrated in Fig. 9 can provide, therefore, a band filter, throughout the transmission band of which the nature of the phase vor delay characteristic, can be precisely controlled to provide any one of' a large number of widely different phase, or delay, versus frequency characteristics, simply by appropriately spacing the zeros and poles of the two groups of resonators in precisely the manner taught by Bode andv Bode and Dietzold in their above-mentioned papers. Phase" or delayequalization over any desired portion of the guided wave frequency range can therefore be realized by the use of such a structure having a transmitting band which includes the portion of the frequency range to be equalized. A larger or smaller number of critical frequencies can of course readily be realized by :increasing or decreasing the number of resonators in each group. In general, the transmission band of the phase or delay 'equalizing structure should substantially overlap bothends of thefrequency range over which precise phase or delay equalization is-desired so that the more abrupt changes 'usually encountered near the cut-01T (band edge) frequencies will not fall within the frequency region being employed. It might here be noted in passing, thatl Bode teaches, for lumped-element low frequency structures, the spacing of critical frequencies to obtain a substantially linear phase characteristic throughout the entire transmission band and even somewhat beyond the cutoff frequencies but such complicated arrangements do not yet appear necessary in the guided wave frequency region. The possibility of obtaining such characteristics, however, is here noted as involving merely a straight-forward application of the principles of the present invention. Where a more steeply rising attenuation characteristic providing greater attenuation over portions of the attenuated frequency regions not far removedfrom the cut-01T' frequencies,` it will at times `be desirable to position Vsome of the critical frequencies of both the direct and contrary -coupled resonators in such attenuated frequency regions. One such an arrangement is illustrated by the admittance diagrams shown in Fig. 42 and will be described in detail hereinunder.

The admittance curves of an illustrative very simple band-pass wavelter of the invention, employing only two direct coupled resonators and one contrary coupled resonator areshown YorY is zero, the third term is zero and I1/I2= +;l. Thus, if we make the poles of coincide with the zeros of and the zeros of coincide with the poles of we will have prefect transmission at the frequencies of these pole-zero combinations.

Further, somewhere between a pole and a zero of we will have For a nearly even spacing of poles and zeros this will be about half-way between a pole and a zero. For the frequency at which Equation 13 holds we have l=ij M/2B+2B/M (14) From this it is obvious that at this frequency we will have perfect transmission with i 90- degree phase shift if Thus: (l) if we tune the directly coupled and contrarily coupled resonators so that the poles of one set lie at the same frequencies as the zeros of the other set, at these frequencies we get perfect transmission with i 180-degree phase shift (2) if We adjust the coupling of the resonators so that at frequencies Where Athe absolute values of Y and Y `are equal to M /2, we will get perfect transmission at these frequencies.

This presents, by a simple process of spot-f check analysis, a, reasonably clear picture of the action of a properly adjusted paralleled resonator transducer. Suppose we consider a frequency within the transmitted band of frequencies at which a direct coupled resonator is resonant. At this frequency the sum of transmission through contrary coupled resonators is zero. As we raise the frequency, a contrary coupled resonator tuned to a higher frequency begins to transmit. About half-way between the resonant frequency of the direct coupled resonator and that of the contrary coupled resonator the transmission between the two is equal and the phase shift is degrees. At a still higher frequency all transmission will be through the contrary coupled resonator and the direct coupled resonators will transmit nothing; the phase will then be degrees. As the frequency is raised, the phase will thus gradually increase and transmission will shift from one resonator to another.

This immediately tells us (in conformance with fundamental filter theory) that to get a substantially linear phase characteristic throughout the transmitted band of frequencies We need merely provide equal frequency spacings between zeros and poles.v If, however, we want to get as sharp a cut-off as possible for a given number of resonators it is then necessary to tune to frequencies which are relatively far apart and to couple tightly the resonators which are resonant near the center of the band and to tune to frequencies which are relatively close together and couple loosely the resonators which are resonant near the edges of the band, always, of course, throughout the transmitted band, making the zeros of one set of resonators (direct coupled) and the poles of the other set (contrary" coupled) coincide, and fulfilling the relations expressed by Equations 14 and 13. This gives a slow variation of phase with frequency near the center of the band and a rapid variation near the edges. The principles involved are precisely those evolved by Bode and Dietzold for low frequency lattice type lumped-element structures. They are explained in the Bode, and Bode and Dietzold, papers mentioned above, and are also summarized by Terman at pages 238 to 244 of his above-mentioned handbook.

Furthermore, precisely as taught by Bode for lumped-element structures, in the frequency regions it is desired to suppress in the output of the filtering structure the zeros, if any, of the directly coupled reactive structures in these regions should correspond in frequency with zeros of the contrarily coupled reactive structures in these regions and poles of the directly coupled structures should correspond in frequency with the poles of the contrarily coupled structures. An odd zero or pole in one or the other group of structures defines the cut-off frequency at the edge of a transmitting region. For example, see Fig. 104 on page 239 of the above-mentioned Radio Engineers Handbook by F. E. Terman. Such arrangements are illustrated by the diagrams of Figs. 42 and 43 and explained in detail hereinunder. It is frequently convenient, but is not necessary, that the resonators, reckoned in order of their successively increasing frequencies of resonance, be distributed along the interconnected transmission lines with direct and contrary coupled resonators occurring alternately.

Branching filters We have, so far, said nothing about branching filters. Suppose we examine Equation 11, with respect to the input admittance of a lter. We see that far outside the band, where i. YandY approach zero the input admittance Yi approaches zero. This means that outside of the band both the conductance Gi and the susceptance Bi, the real and imaginary components of admittance Yi, respectively, must decrease as we go further from the band.

From the symmetry of the admittance plots of Fig. 9 it is almost obvious that for a filter adjusted symmetrically with respect to frequency, at a given distance from the center above the band, Gi will be the same as at a given distance from the center below the band, but the susceptance in the two cases will be equal and opposite. Suppose, then, We couple in tandem two similar filters, for the rst of which Gi=M/2 at some frequency fm above the band and for the second of which Gi=M/2 but fm lies below the band. Then, the sum of the susceptances will be Zero at fm; the sum of the conductances will be M, and the two filters in tandem will provide a match for the guide to which they are coupled. t

As both are matched in their bands, the match through one band and into the other should be good.

Thus, a branching lter as shown in Fig. 4 can be made to present a good match looking into t' the common wave guide over all the bands involved. In this case, however, the bands will overlap somewhat. If we do not want the bands to overlap We can (1) merely separate the bands sufficiently so that they do not overlap seriously, leaving the common guide unmatched in the intervals between the bands, or (2) provide dummy resonators arranged to simulate the input impedances of nlters which pass the inter-band intervals and which work into dissipative or resistive terminations as a load Fig. 10 shows a simple form of the last-mentioned type of branching filter. Here we have a ltwo-way branching lter, with bands B1 and B3 present in wave guide |60 being separated and of..

directed to output wave guides |E| and |97., respectively. The resonators |05 and |06 between them pass band B3, excluding band B1, while resonators lill and IBS between them pass band B1 and exclude band B3. provided to adjust the tuning of these resonators. Energy in an intervening band Bz goes into a resonator ISB with adjustable tuning stub |99, adjustable coupling to the common guide and adjustable loss stub lll. (The loss can be adjusted, for instance, by screwing the stub EM of lossy, i. e. energy dissipating, material further into or out of the resonator, as is well known to those skilled in the art.) The parameters of this resonator are adjusted to provide an impedance match to the common guide in the frequency range between the frequency bands B1 and B3. Adjustable couplings suitable for use with device |03 and resonators ii to lii, inclusive, will be described hereinunder. As an alternative way of viewing the situation we can consider that, in Fig. l0, output B2 is terminated in its characteristic admittance and suffered to exist merely as a means for providing an appropriate im- Tuning stubs H39 are e 18 pedance termination for the common guide |00 between bands B1 and B3.

Specific suggestions for resonator shape, coupling, etc.

As we have seen, some of the resonators in a paralleled resonator transducer preferably provide coupling between input and output guides in one sense, and others in the other or opposite Sense (direct and contrary coupling). There are at. least twoy solutions to this problem leading to somewhat different paralleled resonator transducer designs. One of these is to locate the resonators for either shunt or series relation substantially as shown in Figs. 1 to 4, inclusive, and described in detail above and to provide coupling means which are somewhat dilferent for different resonators. Such coupling means are described hereinafter. The other solution is to provide similar coupling means for all resonators but to locate the resonators somewhat differently. This latter solution will be discussed below.

Direct and contrary coupling means Figs. 11A and 11B to 13, inclusive, illustrate, by way of example, another possible construction for an 8-resonator transducer of the invention. The input and output guides lll] and are overlapped with their larger dimensions contiguous, as shown, and are coupled by four resonators on each side. Wave guide lill is closed at its right end and wave guide II is closed at its left end. In the front side View of Fig. 11B, the four resonators on the front side are designated ||3 to H6, inclusive. Four similar resonators I|3a to Ila, shown in top view Fig. 11A, are assembled on the back side, immediately behind the front resonators, ||3 to I6, respectively. Resonator |6a appears also in the end view of Fig. 12. The eight resonators (front) ||3 to H6, inclusive, and (rear) ||3a to Ha, inclusive, can preferably be of rectangular coniiguration, their respective dimensions as shown in the cross-sectional View of Fig. 13 through resonators I5, |5a being approximately freespace wavelengths in depth of the median frequency of the band to be transmitted through any particular resonator. The coupling means are the wires, I8 to |2l, inclusive, shown in both Figs. 12 4and 13, extending in the direction of the electric elds, |21, |21 and |26, |26 of Fig. 13, in both the resonators and the guides, respectively, as shown more clearly in the cross-sectional View taken through resonators H5, lla and the two guides H0, in Fig. 13. Coupling wires I I8 to |2| can, for example, be of round wire, approximately one-sixteenth inch in diameter and, for example, of a total length of one-half wavelength, or somewhat more, or less, if it is desired to avoid resonant couplings. These coupling wires should have portions parallel to the electric eld in the wave guides, or resonators, which are extended approximately halfway or less from the center `to the nearest wall surface. Arrows |26, |26', |21, |21', of Fig. 13, indicate the directions of the electric vectors in the wave vguide-s and resonators, respectively, as shown.

The coupling wires H8 to |2|, inclusive, can be held centrally with respect to apertures |22 to |25, shown in Fig. 13, inclusive, respectively, by dielectric washers or bearings, not shown. Any

1 9 suitable low loss, low dielectric-constant material such as polystyrene can be employed for this purpose. Rotation of the wires about their horizontal portions .as shown in Figs. 12 and 13 can change the magnitude and phase of the coupling; settings for direct coupling (resonator H5) and contrary coupling (resonator ll5a) are shown in Fig. 13. The positions of the coupling wires IIB to |2I shown in Fig. 12 correspond to a direct coupling of resonator |I6a and a contrary coupling of resonator IIS between the wave guides ||0, Intermediate positions will result in decreased magnitude of coupling, the minimum coupling being obtained when one wire of a pair is at 90 degrees with respect to the other. Individual tuning stubs I2, are provided with each resonator to change the resonant frequencies of the resonators |I3 to |16 and ll3a, to H60., inclusive, as necessary to obtain the characteristics required. The resonators on each side should be coupled to the input and output guides at points substantially one-half wave, of the median frequency of the respective bands transmitted, apart and at an odd number of quarter wavelengths from the closed ends of the wave guides, as for the structures of Figs. 1 to 3, inclusive.

Figs. 14 and 15 illustrate, in cross-sectional views, another physical arrangement for coupling input and output guides by resonators.

In Fig. 14, the input wave guide |33 is coupled to the output wave guide |28 by `the intermediately placed resonator |3I. The directions of the electric vectors are indicated by arrows |34 and |35 for the wave guides |33 and |28 respectively, and by arrow |44 for the resonator !3|. Tuning stub |30 permits adjustment of the resonant frequency of resonator 13| by, in effect, changing the capacity of the resonant circuit. Coupling wires |32 and |28 act as antennas in the wave guides and resonator and can provide an appreciable additional direct -coupling between wave guides |33, |28. Again, the coupling wires can be held by -dielectric washers or bearings (not shown) in the orices between the resonator and the guides and the phase and magnitude of the coupling can be varied by rotating one coupling wire with respect to the other about its vertical portion, with results in general similar to those r obtaining in the arrangement of the tyne illustrated in Fig. 13, and described above. The coupling between the wave guides as shown in Fig. 14 is of the "contrary type.

In Fig. 15 the input wave guide |31 is coupled to the output wave guide |36 by the intermediately placed resonator 14|. Arrows |38 and |39 indicate the directions of the electric vectors for wave guides |31 and |36 respectively. Arrow |45 `indicates the direction of the electric vector in resonator |4| This arrangement provides a direct coupling las contrasted with the contrary coupling as shown in Fig. 14, but is otherwise similar.

Figs. 16 and 17 show two cross-sectional views of a coupling between input and output guides |46 and |41 by an I-I-O mode resonator |53. Such resonators have low loss. The H-O mode resonator is disclosed and described, for example, in the handbook Reference Dat-a for Radio Engineers, published by the Federal Telephone and Radio Corporation, 67 Broad Street, New York, New York, 2nd edition, 1946, at pages 215 and 221. In Figs. 16 and 17 arrows |48 and |49 indicate the direction of the electrical vector in wave guides |41 and |46 respectively. Arrows |54, in Fig. 17, indicate the direction of the electric vector within the resonator |53. Resonator |53 is substantially in the form of a simple right circular cylinder. Its diameter can be at least .725 free-space wavelength, or greater, and its height. or length along the longitudinal axis, at least onehalf wavelength, for H-O propagation in such a cylinder, of the median frequency of the frequency band to be transmitted from one wave guide to the other. The length will thus usually be somewhat more th-an one-half free-space wavelength. The orifices |55 can be circular openings or" approximately f6 or 1/20 wavelength diameter. The wires |50 and 15| act as antennas to couple the resonator |53 to the guides |46 and |41. They can be substantially one-half free-space wavelength long to provide a resonant antenna with a strong coupling, or somewhat shorter, or longer, if non-resonant coupling is dcsired. The coupling will be greater the farther the wire 'extends into the wave guides. Minimum coupling will be attained if the bent portions of these wires yare located about half-way between the wall and the center of the resonator Figs. 18 and 19 show an arrangement of input and output guides and resonators using irises instead of bent wires as the coupling means. In Fig. 18 the input wave guide |13 is coupled to the output wave guide |15 by a resonator |10. The directions of the electric fields in the guides are shown by arrows |14 and |16, respectively, and the direction of the eld in the resonator is shown by the arrow |1I. The resonator can be tuned by the tuning stub |12. The irises |11 and |18 couple the resonator to the guides. Part of the current which ilows in the walls of the guide |13' (assuming it to be the input guide), transverse to the direction of propagation, excites, in the vicinity of the aperture |18, a current in the resonator |10. Part of the wall current of the resonator |10 excites, in the vicinity of the aperture |11, a transverse wall current in the wave guide |15. The excitation of the current, that is, the coupling, will be greater the larger are the apertures.

For operation at 4,000 megacycles, by Way of example, the wave guides |13 and |14 can be one inch high and two inches wide. The resonator |10 .can be two inches high, two inches wide and two inches deep normal to the plane of the paper. The apertures |13 and 11 can be holes about one inch in diameter, or one inch square.

Fig. 19 is similar to Fig. 18. The guides |80, |85, the resonator |82, the arrows |81, |04, |86, the tuning stub |83 and the iris |88 correspond respectively to |13, |15, |10, |14, |1|, |16, |18 of Fig. 18. The iris of |81 of Fig. 19 is differently placed from |11 of Fig, 18 so as to excite a wave in guide |85, opposite in phase to that excited in guide 15, and hence to provide contrary" instead of direct coupling.

These examples. of course, by no means exhaust the general approach illustrated. Numerous and varied other arrangements embodying the principles of the invention and within the scope and spirit thereof, will readily occur t0 those skilled in the art. 

